The present invention relates to a boosted switch device for a sampler of an analog/digital converter, and to an operating method thereof.
As is known, broadly speaking, analog/digital conversion systems can be subdivided into two main categories, according to the conversion principle on the basis of which they operate: the first category includes the so-called Nyquist analog/digital conversion systems, which can be schematized with a continuous-time anti-aliasing filter, a switched-capacitor channel filter, and a Nyquist analog/digital converter, which are connected to one another in cascade, whereas the second category includes the so-called over-sampling analog/digital conversion systems, which can be schematized with a continuous-time anti-aliasing filter, an over-sampling analog/digital converter, and a digital channel filter which are connected to one another in cascade.
However, both categories carry out the conversion of an analog signal, i.e., of a signal which is continuous in time and in amplitude, into a sampled data signal, i.e., which is continuous in time and discrete in amplitude, by means of a so-called sampler.
FIG. 1 shows the general circuit diagram of the sampler which is best known and most commonly used in the applications.
In particular, as illustrated in FIG. 1, the sampler, indicated as a whole by 1, comprises an input terminal 2, at which there is present an input voltage VA to be sampled; an output terminal 4; a sampling capacitor 6 connected between a first and a second node 8, 10; a first switch device 12 controlled by a first control signal F1D, and connected between the input terminal 2 of the sampler 1 and the first node 8; a second switch device 14 controlled by a second control signal F2D, and connected between the first node 8 and a ground line 16 set to a ground potential VGND, typically 0 V; a third switch device 18 controlled by a third control signal F1, and connected between the second node 10 and the ground line 16; and a fourth switch device 20 controlled by a fourth control signal F2, and connected between the output terminal 4 of the sampler 1, and the second node 10.
The output terminal 4 of the sampler is then typically connected to the virtual ground of an operational amplifier (not shown), with which it forms a switched-capacitor integrator.
On the other hand, FIG. 2 shows the temporal development of the four control signals F1, F2, F1D, F2D, which are commonly also known as xe2x80x9cphases,xe2x80x9d and are supplied to the switch devices 12, 14, 18, 20 of the sampler 1. In particular, it can be noted that the first and the second control signals F1D and F2D are exact replicas respectively of the third and fourth control signals, which are temporally delayed compared with the latter, by a delay TR of approximately a few nanoseconds.
In addition, in order to guarantee correct operation of the sampler 1, the first and second control signals F1D and F2D do not overlap one another temporally, in other words the first and second control signals F1D and F2D never assume a high logic level simultaneously, just as the third and fourth control signals F1 and F2 do not overlap one another temporally. In addition, neither the first and fourth control signals F1D, F2, nor the second and third control signals F2D, F1 overlap one another temporally.
The operation of the sampler 1 is known, and will thus be described here briefly and only to the extent necessary for understanding of the problem on which the present invention is based.
In particular, according to the logic levels assumed by the control signals F1, F2, F1D and F2D, the sampler capacitor 6 is connected cyclically and in an alternating manner between the input terminal 2 and the ground line 16, and between the latter and the output terminal 4. In detail, when the first and the fourth control signals F1, F1D assume a high logic level, the input voltage VA is sampled, and the sample is stored in the sampler capacitor 6, whereas when the second and the third control signal F2, F2D assume a high logic level, the sample of the input voltage VA which is stored in the sampler capacitor 6 is transferred to the output terminal 4 of the sampler 1, and consequently to the virtual ground of the operational amplifier, with which it forms the aforementioned switched-capacitor integrator.
The use of four control signals having the timings shown in FIG. 2 has been proposed in xe2x80x9cLow-Distortion Switched-Capacitor Filter Design Techniques,xe2x80x9d Kuang-Lu Lee and Robert G. Mayer, IEEE Journal of Solid-State Circuits, vol. sc-20, No. 6, December 1985, Section III B, pages 1103-1112, in order to overcome the disadvantages of the prior art samplers, in which the first and second switch devices 12, 14 are controlled respectively by the control signal F1 and by the control signal F2, instead of by their temporally delayed replicas.
In particular, the prior art samplers had the disadvantage that they introduced onto the output signal VB unacceptable distortions, generated mainly by the switch devices 12, 14, 18, 20. In fact, since these switch devices are not ideal, but have capacitances, and thus associated charges, which vary strongly according to the input voltage VA, at the instant at which these switch devices open, they introduce onto the output signal VB distortions which detract considerably from the performance of the sampler.
As demonstrated in the aforementioned article, by using instead, the control signals shown in FIG. 2 with the structure shown in FIG. 1, the distortion of the output signal VB is strongly reduced, for input frequencies lower than 10 kHz, i.e., for harmonics of the input voltage VA with frequencies lower than 10 kHz.
However, at high frequencies, and in particular at input frequencies greater than 100 kHz, there is an ever greater increase in other distortions caused mainly by the circuit structure of the first switch device 12, and on which the solution proposed in the aforementioned article does not have any effect, as explained in depth in xe2x80x9cA Cascaded Sigma-Delta Pipeline A/D Converter with 1.25 MHz Signal Bandwidth and 89 dB SNR,xe2x80x9d Todd L. Brooks, David H. Robertoson, Daniel F. Kelly, Anthony Del Muro and Stephen W. Harston, IEEE Journal Solid-State Circuits, vol. 32, No. 12, December 1997, Section IV B, pages 1896-1905.
In particular, according to the prior art, the first switch device 12 was usually formed by a CMOS transfer-gate switch having the circuit structure shown in FIG. 3, i.e., formed by an NMOS transistor 22 and by a PMOS transistor 24 having drain terminals connected to one another and to the input terminal 2 of the sampler 1, source terminals connected to one another and to the first node 8, and gate terminals receiving respectively the first control signal F1D and the first inverted control signal {overscore (F1D)}, the latter being generated by means of a logic inverter (not shown).
As explained in the aforementioned article, the switch devices with the circuit structure shown in FIG. 3 have series resistances that vary considerably according to the input voltage VA, thus causing strong distortions at a high frequency.
In order to overcome the limitations inherent in the switch devices having the circuit structure shown in FIG. 3, in the aforementioned article, a switch device having the circuit structure shown in FIG. 4 is proposed, which, for the reasons given hereinafter, is commonly known as xe2x80x9cbootstrapped clock-boosted switch.xe2x80x9d
In particular, as shown in this Figure, the switch device, indicated as a whole by 12xe2x80x2, comprises an NMOS transistor 30 having a drain terminal connected to the input terminal 2 of the sampler 1, a source terminal connected to the first node 8 of the sampler 1, and a gate terminal connected to a third node 32; a PMOS transistor 34 having a drain terminal connected to the third node 32, source and bulk terminals connected to one another and to a fourth node 36, and a gate terminal receiving the first inverted control signal {overscore (F1D)}; and an NMOS transistor 38 having a drain terminal connected to the third node 32, a source terminal connected to the ground line 16, and a gate terminal receiving the second control signal F2D.
The switch device 12xe2x80x2 further comprises an amplifier 40, and a CMOS transfer-gate switch 42, which are connected to one another in series between the input terminal 2 and the fourth node 36. In particular, the amplifier 40 has a unity gain, whereas the CMOS transfer-gate switch 42 is formed by an NMOS transistor 44 and a PMOS transistor 46, having drain terminals connected to one another and to the output terminal of the amplifier 40, source terminals connected to one another and to the fourth node 36, and gate terminals receiving respectively the second control signal F2D and the second inverted control signal {overscore (F2D)}, the latter being generated by means of a logic inverter (not shown).
The switch device 12xe2x80x2 further comprises a so-called bootstrap capacitor 48 connected between the fourth node 36 and a fifth node 50; a PMOS transistor 52 having a drain terminal connected to the fifth node 50, a source terminal connected to a supply line 54 set to a supply potential VCC, typically approximately 3 V, and a gate terminal receiving the first inverted control signal F1D; and an NMOS transistor 56 having a drain terminal connected to the fifth node 50, a source terminal connected to the ground line 16, and a gate terminal receiving the second control signal F2D.
The operation of the switch device 12xe2x80x2 is described fully in the aforementioned article, and is therefore summarized hereinafter only to the extent necessary for understanding of the problem on which the present invention is based.
In particular, the NMOS transistor 30 acts as a switch, and is on when the first control signal F1D is in the high logic state, and is off when the first control signal F1D is in the low logic state, whereas the bootstrap capacitor 48 acts as a floating battery.
When the control signal F2D is in the high logic state, and therefore, owing to the non-overlapping condition, the first control signal F1D is in the low logic state, the NMOS transistors 38 and 56 are on, the CMOS switch 42 is closed, and the PMOS transistors 34 and 52 are off. Thus, the third node 32 is connected to the ground line 16, and therefore keeps the NMOS transistor 30 off, whereas the bootstrap capacitor 48 is connected between the output terminal of the amplifier 40, at which the input voltage VA is present, since the amplifier 40 has a unity gain, and the ground line 16, and therefore the input voltage VA is applied to the terminals of the bootstrap capacitor 48.
On the other hand, when the first control signal F1D is in the high logic state, and the second control signal F2D is in the low logic state, the NMOS transistors 38 and 56 are off, the CMOS switch 42 is open, and the PMOS transistors 34 and 52 are on, such that the bootstrap capacitor 48 is connected between the supply line 60 and the gate terminal of the NMOS transistor 64.
In particular, the fifth node 50 is connected to the supply line 54, and therefore to the fourth node 36, and consequently, a voltage equivalent to VCC+VA is present at the gate terminal of the NMOS transistor 30.
Therefore, depending on the logic levels assumed by the control signals F2D, {overscore (F2D)} and {overscore (F1D)}, the bootstrap capacitor 48 is connected cyclically and in an alternating manner between the input terminal 2 and the gate terminal of the NMOS transistor 30, and between the latter and the supply line 54. Thus, by this means, the gate-source voltage VGS and the drain-source voltage VDS of the NMOS transistor 30 during conduction, are approximately constant, and equivalent to the supply voltage VCC, independently of the input voltage VA (bootstrapping effect of the gate-source voltage) and the potential of the gate terminal of the NMOS transistor 30 exceeds the supply voltage VCC supplied to the switch device 12xe2x80x2 (boosting effect of the gate terminal), and hence the name xe2x80x9cbootstrapped clock-boosted switchxe2x80x9d of the switch device 12xe2x80x2.
Since the series resistance RON of a MOS device during conduction is, in the first approximation, inversely proportional to the difference between the gate-source voltage VGS and the threshold voltage VTH of the MOS device, if the gate-source voltage VGS is independent from the input voltage VA, the sole cause of distortion is that which is associated with dependence of the threshold voltage VTH on the input voltage VA, which is in any case somewhat small.
A first limitation of the switch device 12xe2x80x2 consists in the fact that the input voltage VA must have a bandwidth which is limited to frequencies which are far smaller than the sampling frequency, otherwise the variations in the input voltage VA from one phase to the next, i.e., from the phase in which the first control signal F1D is in the high logic state and the second control signal F2D is in the low logic state, to the opposite phase, in which the first control signal F1D is in the low logic state, and the second control signal F2D is in the high logic state, would be high, the gate-source voltage VGS of the NMOS transistor 30 would not be constant and equivalent to the supply voltage VCC, and therefore all the advantages of the structure described in the aforementioned article would be lost.
This consequently involves the need to use a sampling frequency which is far greater than the Nyquist sampling frequency (it should be remembered that according to the known Nyquist theorem, in order to avoid losing information during the sampling, it is sufficient to use a sampling frequency which is equivalent to double the maximum input frequency, i.e., which is equivalent to double the maximum frequency of the harmonics of the input signal VA), with a consequent waste of power and area on the silicon.
A second limitation of the switch device 12xe2x80x2 consists in the fact that when the first control signal F1D is in the high logic state, the third and fourth nodes 32, 36 go to a voltage which is equivalent to VCC+VA, such that problems of reliability could arise if VCC+VA were greater than the maximum operating voltage which can be withstood by integrated devices, which is determined by the technological process used for manufacture of the integrated devices.
For example, in a technological process at 0.5 xcexcm, the maximum operating voltage which can be withstood by integrated devices is equivalent to 4.6 V, and if the supply voltage VCC were equivalent to 3.3 V, as is typically required by the user, in order for VCC+VA not to exceed the maximum operating voltage, it is necessary to limit the maximum input voltage VA to 1.3 V, and this leads to a loss of signal/noise ratio of the A/D converter in which the switch device 12xe2x80x2 is inserted.
According to an embodiment of the present invention, a boosted switch device for a sampler of an analog/digital converter is provided. The boosted switch device comprises an input terminal and an output terminal; a supply line set to a supply potential; a ground line set to a ground potential; a switch connected between the input and output terminals; a capacitor; and a switch device connecting the capacitor between the supply line and the ground line, when the switch is open, and between the input terminal and the control terminal of the switch, when the switch is closed.
According to another embodiment of the present invention, a sampler for an analog/digital converter is also provided. The sampler of the invention includes a boosted switch input terminal, a boosted switch output terminal, a first switch arranged between the input and output terminals, a charge storage device; and controlled connection means connecting the charge storage device in an alternating manner between a voltage supply line and circuit ground, and between the boosted switch input terminal and a control terminal of the first switch. The sampler may also include a second charge storage device connected to the boosted switch output terminal.
According to another embodiment of the present invention, an operating method of a boosted switch device for a sampler of an analog/digital converter is also provided. The method includes connecting a charge storage means in an alternating manner between a first line set to a first potential and a second line set to a second potential, and between an input terminal of a switch means and a control terminal of the switch means.